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MIC2208
3mmx3mm 1MHz 3A PWM Buck Regulator
General Description
The Micrel MIC2208 is a high efficiency PWM buck (step-down) regulator that provides up to 3A of output current. The MIC2208 operates at 1MHz and has external compensation that allows a closed loop bandwidth of over 100KHz. The low on-resistance internal p-channel MOSFET of the MIC2208 allows efficiencies over 94%, reduces external component count and eliminates the need for an expensive current sense resistor. The MIC2208 operates from 2.7V to 5.5V input and the output can be adjusted down to 1V. The devices can operate with a maximum duty cycle of 100% for use in low-dropout conditions. The MIC2208 is available in the exposed pad 3mm x 3mm MLF-12L package with a junction operating range from -40C to +125C.
Features
* * * * * * * * * * * * * * * * * * * 2.7 to 5.5V supply voltage 1MHz PWM mode Output current to 3A >90% efficiency Adjustable output voltage option down to 1V Ultra-fast transient response External Compensation Stable with a wide range of output capacitance Fully integrated 5A MOSFET switch Micropower shutdown Thermal shutdown and current limit protection Pb-free 3mm x 3mm MLF-10L package -40C to +125C junction temperature range Internal soft-start 5V or 3.3V Point of Load Conversion Telecom/Networking Equipment Set Top Boxes Storage Equipment Video Cards
Applications
Typical Application
MIC2208
3A 1MHz Buck Regulator
MLF and MicroLeadFrame are trademarks of Amkor Technology Micrel, Inc * 2180 Fortune Drive * San Jose, Ca 95131 * USA * tel +1 (408) 944-0800 * fax +1 (408) 474-1000 * http://www.micrel.com
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MIC2208
Ordering Information
Part Number MIC2208YML
.
Output Voltage(1) Adj.
Junction Temp. Range -40 to +125C
Package 3mmx3mm MLF-10L
Lead Finish Pb-free
Pin Configuration
SW 1 VIN 2 PGND 3 SGND 4 BIAS 5 FB 6 EP 12 SW 11 VIN 10 PGND 9 PGOOD 8 EN 7 COMP
3mm x 3mm MLF-12 (ML)
Pin Description
Pin Number
1,12 2,11
Pin Name
SW VIN
Pin Function
Switch (Output): Internal power P-Channel MOSFET output switch Supply Voltage (Input): Supply voltage for the source of the internal P-channel MOSFET and driver. Requires bypass capacitor to GND. Power Ground. Provides the ground return path for the high-side drive current. Signal Ground. Provides return path for control circuitry and internal reference. Internal circuit bias supply. Must be bypassed with a 0.1uF ceramic capacitor to SGND. Feedback. Input to the error amplifier, connect to the external resistor divider network to set the output voltage. Compensation. This is the internal error amplifier output. Connect external compensation components for type II or type III compensation. Enable (Input). Logic level low will shutdown the device, reducing the current draw to less than 5uA. Power Good. Open drain output that is pulled to ground when the output voltage is within +/- 7.5% of the set regulation voltage Connect to ground.
3,10 4 5 6 7 8 9 EP
PGND SGND BIAS FB COMP EN PGOOD GND
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MIC2208
Absolute Maximum Ratings(1)
Supply Voltage (VIN) ...............................-0.3V to +6V Output Switch Voltage (VSW) .....................-1V to +6V Output Switch Current (ISW) ................................. 10A Logic Input Voltage (VEN)......................... -0.3V to VIN Storage Temperature (Ts)................ -60C to +150C ESD Rating(3) ............................................2KV (HBM)
Operating Ratings(2)
Supply Voltage (VIN)............................+2.7V to +5.5V Logic Input Voltage (VEN,VLOWQ) .................. 0V to VIN Junction Temperature (TJ) .............. -40C to +125C Junction Thermal Resistance 3mmx3mm MLF-12L (JA) ....................... 60C/W
Electrical Characteristics (4)
VIN = VEN = 3.6V; L = 1H; COUT = 4.7F; TA = 25C, unless noted. Bold values indicate -40C< TJ < +125C
Parameter
Supply Voltage Range Under-Voltage Lockout Threshold UVLO Hysteresis Quiescent Current Shutdown Current [Adjustable] Feedback Voltage FB pin input current Current Limit in PWM Mode Output Voltage Line Regulation Output Voltage Load Regulation PWM Switch ONResistance Oscillator Frequency Enable Threshold Enable Input Current Soft Start Time Over-Temperature Shutdown over-Temperature Hysteresis Power Good Range Power Good Resistance
Notes:
Condition
(turn-on)
Min 2.7 2.45
Typ
2.55 100
Max 5.5 2.65
Units
V V mV
VFB = 0.9 * VNOM (not switching) VEN = 0V
720 0.1 0.99 0.98 1 1
950
5 1.01 1.02
A A V nA A %
1% 2% (over temperature)
VFB = 0.9 * VNOM VOUT > 2.2V; VIN = VOUT+500mV to 5.5V; ILOAD= 20mA VOUT < 2.2V; VIN = 2.7V to 5.5V; ILOAD= 20mA 20mA < ILOAD < 3A ISW = 50mA VFB = 0.7VFB_NOM (High Side Switch)
100
10
8 0.13 0.2 95
1
200
%
300 0.9 0.5
VOUT =10% to VOUT=90% 1 0.85 0.1 450 160 20 7 IPGOOD 145 10
m MHz V A s
1.1 1.3 2
C C
%
200
1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 3. Devices are ESD sensitive. Handling precautions recommended. Human body model: 1.5k in series with 100pF. 4. Specification for packaged product only.
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Typical Characteristics
3.3V
96 94 EFFICIENCY (%) 92 90 88 86 84 82 80 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A) 3 5VIN 5.5VIN
OUT
MIC2208 Efficiency
4.5VIN EFFICIENCY (%) 100 98 96 94 92 90 88 86 84 82 80 0
2.5V
OUT
MIC2207 Efficiency
94 92 EFFICIENCY (%) 90 88 86 84 82
2.5V
OUT
MIC2207 Efficiency
IN
4.5V
3.3VIN
3.6V
IN
5VIN
5.5VIN
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
80 0
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
1.8V
95 93 EFFICIENCY (%) 91 89 87 85 83 81 79 77 75 0 3V
OUT
MIC2208 Efficiency
90 88 EFFICIENCY (%) 86 84 82 80 78 76 74 72 70 0
1.8V
OUT
IN
MIC2208 Efficiency
95 90 EFFICIENCY (%) 85 80 75 70 0
1.5V
OUT
MIC2208 Efficiency
4.5V
IN
3VIN 3.3VIN
5VIN
3.3V
IN
3.6V
IN
3.6VIN
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
1.5V
85 83 EFFICIENCY (%) 81 79 77 75 73 71 69 67 65 0 5V
OUT
MIC2208 Efficiency
4.5VIN EFFICIENCY (%) 85 83 81 79 77 75 73 71 69 67 65 0
1.2V
OUT
MIC2208 Efficiency
85 83 EFFICIENCY (%) 81 79 77 75 73 71 69 67 65 0
1.2V
OUT
MIC2208 Efficiency
3VIN 3.3VIN
4.5VIN
IN
5.5VIN
3.6V
IN
5VIN
5.5V
IN
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
Load Regulation
1.010 1.008 OUTPUT VOLTAGE (V) 1.006 1.004 1.002 1.000 0.998 0.996 0.994 0.992 3.3V IN 0.990 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
1.0100 FEEDBACK VOLTAGE (V) 1.0080 1.0060 1.0040 1.0020 1.0000 0.9980 0.9960
Feedback Voltage vs. Temperature
1.20 FEEDBACK VOLTAGE (V) 1.15 1.10 1.05 1.00 0.95 0.90
Feedback Voltage vs. Temperature
0
20
40
60
80
0
20
40
60
100
120
80
100
3
TEMPERATURE (C)
TEMPERATURE (C)
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120
-40
-20
-40
-20
0.9940 0.9920 3.3V IN 0.9900
0.85 3.3V IN 0.80
Micrel, Inc.
MIC2208
1.2 FEEDBACK VOLTAGE (V) 1.0 0.8 0.6 0.4 0.2 0 0
Feedback Voltage vs. Supply Voltage
700 QUIESCENT CURRENT (A) 600 500 400 300 200 100 0 0
Quiescent Current vs. Supply Voltage
P-CHANNEL RDSON (mOhms)
120 115 110 105 100 95 90 85 80 75 70 2.7
RDSON vs. Supply Voltage
1 2 3 4 5 SUPPLY VOLTAGE (V)
1 2 3 4 5 SUPPLY VOLTAGE (V)
3.2 3.7 4.2 4.7 5.2 SUPPLY VOLTAGE (V)
160 P-CHANNEL RDSON (mOhms) 140 120 100 80 60 40 20 -40 0
RDSON vs. Temperature
1.2 ENABLE THRESHOLD (V) 1
Enable Threshold vs. Supply Voltage
1.2 ENABLE THRESHOLD (V) 1
Enable Threshold vs. Temperature
0.8 0.6 0.4 0.2 0 2.7 3.2 3.7 4.2 4.7 SUPPLY VOLTAGE (V)
0.8 0.6 0.4 0.2 0 20 40 60 80 100 120 -40 -20 0
3.3VIN 0 20 40 60 80 100 TEMPERATURE (C) 120 -20
TEMPERATURE (C)
3.5 MAX. OUTPUT CURRENT (A) 3
Max. Continuous Current vs. Ambient Temp 3.3V OUT*
*Using recommended layout (1oz copper) and B.O.M.
3.5 MAX. OUTPUT CURRENT (A) 3
Max. Continuous Current vs. Ambient Temp 2.5V OUT*
*Using recommended layout (1oz copper) and B.O.M.
3.5 MAX. OUTPUT CURRENT (A) 3
Max. Continuous Current vs. Ambient Temp 1.8V OUT*
2.5 2
2.5 2 3.3VIN
2.5 2 3.3VIN 5V
5V
1.5 1 5VIN
1.5 1
IN
1.5 1 *Using
IN
0.5
0.5 0 60 70 80 90 100 110 120 AMBIENT TEMPERATURE (C)
0.5 layout (1oz copper) 0 60 70 80 90 100 110 120 AMBIENT TEMPERATURE (C)
and B.O.M.
recommended
0 60 70 80 90 100 110 120 AMBIENT TEMPERATURE (C)
3.5 MAX. OUTPUT CURRENT (A) 3
Max. Continuous Current vs. Ambient Temp 1.0V OUT*
2.5 2 3.3VIN 5V
1.5 1 *Using
IN
0.5 layout (1oz copper) 0 60 70 80 90 100 110 120 AMBIENT TEMPERATURE (C)
and B.O.M.
recommended
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MIC2208
Functional Diagram
VIN VIN P-Channel Current Limit BIAS
HSD
PWM Control
SW SW
EN
Enable and Control Logic
Bias, UVLO, Thermal Shutdown
COMP
Soft Start
EA
FB
1.0V
PGOOD 1.0V
SGND
PGND
MIC2208 Block Diagram
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Functional Characteristics
Continuous Operation
SWITCH VO LTAGE (2V/div.) SWITCH VO LTAGE (2V/div.)
Discontinuous Operation
0A INDUCTOR CURREN T (500mA/di .) v
0A INDUCTOR CURREN T (200mA/di .) v TIME (200ns/di v.)
VIN = 3.3V VOUT = 1V@0.15A
VIN = 3.3V VOUT = 1V@1A
TIME (200ns/di v.)
OUTPUT VOLTAGE (100mV/di .) v
0A OUTPUT CURREN T (2A/div.)
OUTPU VOLTAGE T ENABLE (1V/div.)
INDUCTOR CURREN T (2A/div.)
INPUT CURREN T (1A/div.)
Transient Respons e
Enable Response
VIN = 3.3V VOUT = 1V@3A
VIN = 3.3V. 1V, 3A to 200m A TIME (100s/di v.)
(5V/div.)
TIME (100s/di v.)
Power Good
POWER GOOD (5V/div.) OUTPU VOLTAGE T (1V/div.) POWER GOOD (5V/div.) OUTPU VOLTAGE T (1V/div.)
Power Good
VIN = 3.3V
VIN = 3.3V
ENABLE (5V/div.)
TIME (100s/di v.)
ENABLE (5V/div.)
TIME (40s/di v.)
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Pin Description
VIN Two pins for VIN provide power to the source of the internal P-channel MOSFET along with the current limiting sensing. The VIN operating voltage range is from 2.7V to 5.5V. Due to the high switching speeds, a 10F capacitor is recommended close to VIN and the power ground (PGND) for each pin for bypassing. Please refer to layout recommendations. BIAS The bias (BIAS) provides power to the internal reference and control sections of the MIC2208. A 10 Ohm resistor from VIN to BIAS and a 0.1uF from BIAS to SGND is required for clean operation. EN The enable pin provides a logic level control of the output. In the off state, supply current of the device is greatly reduced (typically <1A). Do not drive the enable pin above the supply voltage. FB The feedback pin (FB) provides the control path to control the output. For adjustable versions, a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. The output voltage is calculated as follows:
R1 VOUT = VREF x + 1 R2
external components. Refer to the compensation section of the datasheet for determining nessasary component values. SW The switch (SW) pin connects directly to the inductor and provides the switching current nessasary to operate in PWM mode. Due to the high speed switching on this pin, the switch node should be routed away from sensitive nodes. This pin also connects to the cathode of the free-wheeling diode. PGOOD Power good is an open drain pull down that indicates when the output voltage has reached regulation. For a power good low, the output voltage is within +/- 10% of the set regulation voltage. For output voltages greater or less than 10%, the PGOOD pin is high. This should be connected to the input supply through a pull up resistor. A delay can be added by placing a capacitor from PGOOD to ground. PGND Power ground (PGND) is the ground path for the MOSFET drive current. The current loop for the power ground should be as small as possible and separate from the Analog ground (AGND) loop. Refer to the layout considerations fro more details. SGND Signal ground (SGND) is the ground path for the biasing and control circuitry. The current loop for the signal ground should be separate from the power ground (PGND) loop. Refer to the layout considerations for more details.
where VREF is equal to 1.0V. COMP The COMP pin is the output of the internal error amplifier. This pin is used to compensate the MIC2208 for stability over a varying range of
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Applications Information
The MIC2208 is a 3A PWM non-synchronous buck regulator. By switching an input voltage supply, and filtering the switched voltage through an Inductor and capacitor, a regulated DC voltage is obtained. Figure 1 shows a simplified example of a non-synchronous buck converter.
Figure 2. Continuous Operation
The output voltage is regulated by pulse width modulating (PWM) the switch voltage to the average required output voltage. The switching can be broken up into two cycles; On and Off. During the on-time, the high side switch is turned on, current flows from the input supply through the inductor and to the output. The inductor current is
Figure 1.
For a non-synchronous buck converter, there are two modes of operation; continuous and discontinuous. Continuous or discontinuous refer to the inductor current. If current is continuously flowing through the inductor throughout the switching cycle, it is in continuous operation. If the inductor current drops to zero during the off time, it is in discontinuous operation. Critically continuous is the point where any decrease in output current will cause it to enter discontinuous operation. The critically continuous load current can be calculated as follows;
2 V VOUT - OUT VIN IOUT = 1MHz x 2 x L Continuous or discontinuous operation determines how we calculate peak inductor current.
Figure 3. On-Time
Continuous Operation
Figure 2 illustrates the switch voltage and inductor current during continuous operation.
charged at the rate;
(VIN - VOUT )
L To determine the total on-time, or time at which the inductor charges, the duty cycle needs to be calculated. The duty cycle can be calculated as; D= VOUT VIN D 1MHz
and the On time is; TON =
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Therefore, peak to peak ripple current is;
1MHz x L Since the average peak-to-peak current is equal to the load current. The actual peak (or highest current the inductor will see in a steady state condition) is equal to the output current plus half the peak-topeak current. For those of you who have not had enough formulas already, here it is;
Ipk -pk =
(VIN - VOUT ) x VOUT
VIN
+ VD ) L The total off time can be calculated as; - TOFF = 1- D 1MHz
VIN 2 x 1MHz x L Figure 4 demonstrates the off-time. During the offtime, the high-side internal P-channel MOSFET turns off. Since the current in the inductor has to discharge, the current flows through the freewheeling Schottky diode to the output. In this case, the inductor discharge rate is (where VD is the diode forward voltage); Ipk = IOUT +
(VIN - VOUT ) x VOUT
Figure 5. Discontinuous Operation
(VOUT
When the inductor current (IL) has completely discharged, the voltage on the switch node rings at the frequency determined by the parasitic capacitance and the inductor value. In Figure 5, it is drawn as a DC voltage, but to see actual operation (with ringing and all) refer to the functional characteristics. Discontinuous mode of operation has the advantage over full PWM in that at light loads, the MIC2208 will skip pulses as nessasary, reducing gate drive losses, drastically improving light load efficiency.
Efficiency Considerations
Calculating the efficiency is as simple as measuring power out and dividing it by the power in; P Efficiency = OUT x 100 PIN Where input power (PIN) is; PIN = VIN x IIN and output power (POUT) is calculated as; POUT = VOUT x IOUT The Efficiency of the MIC2208 is determined by several factors. * *
Figure 4. Off-Time
Rdson (Internal P-channel Resistance) Diode conduction losses Inductor Conduction losses
*
Discontinuous Operation
Discontinuous operation is when the inductor current discharges to zero during the off cycle. Figure 5. demonstrates the switch voltage and inductor currents during discontinuous operation. September 2005 10
* Switching losses Rdson losses are caused by the current flowing through the high side P-channel MOSFET. The amount of power loss can be approximated by;
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PSW = R DSON x IOUT 2 x D
PD = VF x IOUT x (1 - D) For this reason, the Schottky diode is the rectifier of choice. Using the lowest forward voltage drop will help reduce diode conduction losses, and improve efficiency. Duty cycle, or the ratio of output voltage-to-input voltage, determines whether the dominant factor in conduction losses will be the internal MOSFET or the Schottky diode. Higher duty cycles place the power losses on the high side switch, and lower duty cycles place the power losses on the schottky diode. Inductor conduction losses (PL) can be calculated by multiplying the DC resistance (DCR) times the square of the output current;
PL = DCR x IOUT
2
Where D is the duty cycle. Since the MIC2208 uses an internal P-channel MOSFET, Rdson losses are inversely proportional to supply voltage. Higher supply voltage yields a higher gate to source voltage, reducing the Rdson, reducing the MOSFET conduction losses. A graph showing typical Rdson vs input supply voltage can be found in the typical characteristics section of this datasheet. Diode conduction losses occur due to the forward voltage drop (VF) and the output current. Diode power losses can be approximated as follows:
Figure 6. Switching Transition Losses
Normally, when the switch is on, the voltage across the switch is low (virtually zero) and the current through the switch is high. This equates to low power dissipation. When the switch is off, voltage across the switch is high and the current is zero, again with power dissipation being low. During the transitions, the voltage across the switch (VS-D) and the current through the switch (IS-D) are at middle, causing the transition to be the highest instantaneous power point. During continuous mode, these losses are the highest. Also, with higher load currents, these losses are higher. For discontinuous operation, the transition losses only occur during the "off" transition since the "on" transitions there is no current flow through the inductor.
Component Selection
Input Capacitor
A 10F ceramic is recommended on each VIN pin for bypassing. X5R or X7R dielectrics are recommended for the input capacitor. Y5V dielectrics lose most of their capacitance over temperature and are therefore, not recommended. Also, tantalum and electrolytic capacitors alone are not recommended due their reduced RMS current handling, reliability, and ESR increases. An additional 0.1F is recommended close to the VIN and PGND pins for high frequency filtering. Smaller case size capacitors are recommended due to their lower ESR and ESL. Please refer to layout recommendations for proper layout of the input capacitor.
Also, be aware that there are additional core losses associated with switching current in an inductor. Since most inductor manufacturers do not give data on the type of material used, approximating core losses becomes very difficult, so verify inductor temperature rise. Switching losses occur twice each cycle, when the switch turns on and when the switch turns off. This is caused by a non-ideal world where switching transitions are not instantaneous, and neither are currents. Figure 6 demonstrates (Or exaggerates.) how switching losses due to the transitions dissipate power in the switch.
Inductor Selection
The MIC2208 is designed for use with a 1H inductor. Proper selection should ensure the inductor can handle the maximum average and peak currents required by the load. Maximum current ratings of the inductor are generally given in two methods; permissible DC current and saturation current. Permissible DC current can be rated either September 2005 11
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for a 40C temperature rise or a 10% to 20% loss in inductance. Ensure the inductor selected can handle the maximum operating current. When saturation current is specified, make sure that there is enough margin that the peak current will not saturate the inductor.
Dominant Pole
Diode Selection Since the MIC2208 is non-synchronous, a freewheeling diode is required for proper operation. A schottky diode is recommended due to the low forward voltage drop and their fast reverse recovery time. The diode should be rated to be able to handle the average output current. Also, the reverse voltage rating of the diode should exceed the maximum input voltage. The lower the forward voltage drop of the diode the better the efficiency. Please refer to the layout recommendations to minimize switching noise. Feedback Resistors The feedback resistor set the output voltage by dividing down the output and sending it to the feedback pin. The feedback voltage is 1.0V. Calculating the set output voltage is as follows;
R1 VOUT = VFB + 1 R2
Gain (dB)
Zero 20dB/Decade LC Frequency
Compensation
The MIC2208 utilizes voltage mode compensation and has the error amplifier pin (COMP) pinned out to allow it to be compensated using external components. This allows the MIC2208 to be stable with a wide range of inductor and capacitor values.
Where R1 is the resistor from VOUT to FB and R2 is the resistor from FB to GND. The recommended feedback resistor values for common output voltages is available in the bill of materials on page x. Although the range of resistance for the FB resistors is very wide, R1 is recommended to be 10K. This minimizes the effect the parasitic capacitance of the FB node.
TYPE II compensation Type II compensation can be expressed as polezero-pole. In our case, a dominant pole (R1 and C3) followed by a zero (C3 and R4) , allowing the final pole to be provided by the output inductor and output capacitor (L and COUT). This mode of compensation works well when using higher ESR output capacitors, such as tantalum and electrolytic dielectrics. The ESR of the capacitor, along with the output capacitance provides a zero (COUT and ESR) that negates one of the two poles created by the inductor-output capacitor filter. This allows the gain to cross the 0dB point with a -1 slope (-20dB/decade).
Bias filter A small 10 Ohm resistor is recommended from the input supply to the bias pin along with a small 0.1uF ceramic capacitor from bias-to-ground. This will bypass the high frequency noise generated by the violent switching of high currents from reaching the internal reference and control circuitry. Tantalum and electrolytic capacitors are not recommended for the bias, these types of capacitors lose their ability to filter at high frequencies.
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Type II Compensation
80 70 60 50 40 Gain (dB) 30 20 10 0 -10 -20 100 VIN=5VIN VOUT=1.0V IOUT=3A 288 252 216 180 144 108 72 36 0 -36 1k 10k 100k Frequency (KHz) -72 1M
MIC2208
Bill of Materials
Item Part Number C1a,C1b C2012JB0J106K
08056D106MAT C2 C3 C4 D1 L1 0402ZD104MAT 0402ZD100MAT TPME477M010R0030 SSA33L RLF7030-1R0N6R4 744 778 9001 IHLP2525AH-01 1 R1 R2 CRCW04023012F CRCW04022002F CRCW04023742F CRCW04026042F CRCW04021503F R4 R5 R6 U1
Description
10uF Ceramic Capacitor X5R 0805 6.3V 10uF Ceramic Capacitor X5R 0805 6.3V 0.1uF Ceramic Capacitor X5R 0402 10V 100pF Ceramic Capacitor X5R 0402 10V 470uF Tantalum Capacitor 10V 3A Schottky 30V SMA 1uH Inductor 8.8mOhm 7.1mm(L) x 6.8mm (W)x 3.2mm(H) 1uH Inductor 12mOhm 7.3mm(L)x7.3mm(W)x3.2mm(H) 1uH Inductor 17.5m (L)6.47mmx(W)6.86mmx(H) 1.8mm 30.1K 1% 0402 resistor 20 k 1% 0402 For 2.5VOUT 37.4 k 1% 0402 For 1.8 VOUT 60.4 k 1% 0402 For 1.5 VOUT 150 k 1% 0402 For 1.2 VOUT Open For 1.0 VOUT 499K 1% 0402 resistor 10 1% 0402 resistor 10K 1% 0402 resistor 1MHz 3A Buck Regulator
Manufacturer
TDK Murata AVX AVX AVX AVX Vishay Semi TDK Wurth Electronik Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Micrel
Qty
GRM219R60J106KE19 10uF Ceramic Capacitor X5R 0805 6.3V
2 1 1 1 1
1 1
1 1 1 1 1
CRCW04024993F CRCW040210R0F CRCW04021002F MIC2208BML
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TYPE III compensation Type III in our case, is a dominant pole (C3 and R1) followed by a zero (C3 and R4) and an additional zero (C5 and R4), allowing the final pole to be provided by the output inductor and output capacitor. This mode of compensation is required when using low ESR output capacitors, such as ceramic capacitors. The additional zero offsets the double pole created by the inductor/output capacitor filter.
Type III Open Loop Gain Response
Dominant Pole
Gain(dB)
Zero Zero
LC Frequency 20dB/Decade
Frequency (Hz)
Type III Compensation
80 70 60 50 40
288 V IN=5V IN V OUT=1.0V IOUT=3A COUT=47F 252 216 180 144 108 72 36 Gain Phase 0 -36 100 1k 10k 100k -72 1M
Gain (dB)
30 20 10
MIC2208
-10 -20
Frequency (KHz)
Bill of Materials
Item
C1a,C1b C2 C3 C4 C5 D1 L1
Part Number
C2012JB0J106K GRM219R60J106KE19 08056D106MAT 0402ZD104MAT 0402ZD103MAT C3216X5R0J476K 12106D476MAT2A VJ0402A330KXAA SSA33L RLF7030-1R0N6R4 744 778 9001 IHLP2525AH-01 1
Description
10uF Ceramic Capacitor X5R 0805 6.3V 10uF Ceramic Capacitor X5R 0805 6.3V 10uF Ceramic Capacitor X5R 0805 6.3V 0.1uF Ceramic Capacitor X5R 0402 10V 1nF Ceramic Capacitor X5R 0402 10V 47uF Ceramic Capacitor X5R 1206 6.3V 47uF Ceramic Capacitor X5R 1210 6.3V 33pF Ceramic Capacitor 0402 3A Schottky 30V SMA 1uH Inductor 8.8mOhm 7.1mm(L) x 6.8mm (W)x 3.2mm(H) 1uH Inductor 12mOhm 7.3mm(L)x7.3mm(W)x3.2mm(H) 1uH Inductor 17.5m (L)6.47mmx(W)6.86mmx(H) 1.8mm 49.9K 1% 0402 resistor
Manufacturer
TDK Murata AVX AVX AVX TDK Murata AVX Vishay VT Vishay Semi TDK Wurth Electronik Vishay Dale Vishay Dale
Qty
2 1 2 1 1 1
GRM32ER60J476ME20 47uF Ceramic Capacitor X5R 1206 6.3V
1 1
R1
CRCW04024992F
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Micrel, Inc.
MIC2208
Bill of Materials (cont.)
CRCW04023322F CRCW04026192F R2 CRCW04021003F CRCW04022493F R3 R4 R5 R6 U1 CRCW04024991F CRCW04024991F CRCW040210R0F CRCW04021002F 33.2 k 1% 0402 For 2.5VOUT 61.9 k 1% 0402 For 1.8 VOUT 100 k 1% 0402 For 1.5 VOUT 249 k 1% 0402 For 1.2 VOUT Open For 1.0 VOUT 4.99K 1% 0402 resistor 90.9K 1% 0402 resistor 10 1% 0402 resistor 10K 1% 0402 resistor 1MHz 3A Buck Regulator Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale Vishay Dale 1 1 1 1 1 1
MIC2208BML
Micrel
Loop Stability and Bode Analysis
Bode analysis is an excellent way to measure small signal stability and loop response in power supply designs. Bode analysis monitors gain and phase of a control loop. This is done by breaking the feedback loop and injecting a signal into the feedback node and comparing the injected signal to the output signal of the control loop. This will require a network analyzer to sweep the frequency and compare the injected signal to the output signal. The most common method of injection is the use of transformer. Figure 7 demonstrates how a transformer is used to inject a signal into the feedback network.
be able to inject high frequencies. Transformers with these wide frequency ranges generally need to be custom made and are extremely expensive (usually in the tune of several hundred dollars!). By using an op-amp, cost and frequency limitations used by an injection transformer are completely eliminated. Figure 8 demonstrates using an op-amp in a summing amplifier configuration for signal injection.
Network Analyzer "R" Input Feedback Network Analyzer "A" Input Output
+8V
MIC922BC5
R1 1k
R3 1k
R4 1k 50
Network Analyzer Source
Figure 8. Op Amp Injection
Figure 7. Transformer Injection
A 50 ohm resistor allows impedance matching from the network analyzer source. This method allows the DC loop to maintain regulation and allow the network analyzer to insert an AC signal on top of the DC voltage. The network analyzer will then sweep the source while monitoring A and R for an A/R measurement. While this is the most common method for measuring the gain and phase of a power supply, it does have significant limitations. First, to measure low frequency gain and phase, the transformer needs to be high in inductance. This makes frequencies <100Hz require an extremely large and expensive transformer. Conversely, it must September 2005 15
R1 and R2 reduce the DC voltage from the output to the non-inverting input by half. The network analyzer is generally a 50 Ohm source. R1 and R2 also divide the AC signal sourced by the network analyzer by half. These two signals are "summed" together at half of their original input. The output is then gained up by 2 by R3 and R4 (the 50 Ohm is to balance the network analyzer's source impedance) and sent to the feedback signal. This essentially breaks the loop and injects the AC signal on top of the DC output voltage and sends it to the feedback. By monitoring the feedback "R" and output "A", gain and phase are measured. This method has no minimum frequency. Ensure that the bandwidth of the op-amp being used is much greater than the expected bandwidth of the power supplies control loop. An op-amp with
M9999-092905 www.micrel.com
Micrel, Inc.
MIC2208
>100MHz bandwidth is more than sufficient for most power supplies (which includes both linear and switching) and are more common and significantly cheaper than the injection transformers previously mentioned. The one disadvantage to using the opamp injection method, that the supply voltages need to below the maximum operating voltage of the opamp. Also, the maximum output voltage for driving 50 Ohm inputs using the MIC922 is 3V. For measuring higher output voltages, a 1MOhm input impedance is required for the A and R channels. Remember to always measure the output voltage with an oscilloscope to ensure the measurement is working properly. You should see a single sweeping sinusoidal waveform without distortion on the output. If there is distortion of the sinusoid, reduce the amplitude of the source signal. You could be overdriving the feedback causing a large signal response.
By setting up a network analyzer to sweep the feedback current, while monitoring the output of the voltage regulator and the voltage across the load resistance, output impedance is easily obtainable. To keep the current from being too high, a DC offset needs to be applied to the network analyzer's source signal. This can be done with an external supply and 50 Ohm resistor. Make sure that the currents are verified with an oscilloscope first, to ensure the integrity of the signal measurement. It is always a good idea to monitor the A and R measurements with a scope while you are sweeping it. To convert the network analyzer data from dBm to something more useful (such as peak-to-peak voltage and current in our case);
V =
dBm 10 10
x 1mW x 50 x 2 0.707
Output Impedance and Transient response
Output impedance, simply stated, is the amount of output voltage deviation vs. the load current deviation. The lower the output impedance, the better. Z OUT = VOUT IOUT
and peak to peak current;
dBm 10 10
I =
x 1mW x 50 x 2 0.707 x R LOAD
The following graph shows output impedance vs frequency at 2A load current sweeping the AC current from 10Hz to 10MHz, at 1A peak to peak amplitude.
Output impedance for a buck regulator is the parallel impedance of the output capacitor and the MOSFET and inductor divided by the gain;
Z TOTAL = R DSON + DCR + X L X COUT GAIN
Output Impedance (Ohms)
Output Impedance vs Frequency
1
To measure output impedance vs. frequency, the load current must be load current must be swept across the frequencies measured, while the output voltage is monitored. Figure 9 shows a test set-up to measure output impedance from 10Hz to 1MHz using the MIC5190 high speed controller.
3.3V IN 0. 1
5V IN 0 . 01 V OUT = 1.8V L =1H COUT = 4.7F+0.1F
0. 0 0 1 10
10 0
1k
10 k
10 0 k
1M
10 M
Frequency (Hz)
Figure 9. Output Impedance Measurement
From this graph, you can see the effects of bandwidth and output capacitance. For frequencies <100KHz, the output impedance is dominated by the gain and inductance. For frequencies >100KHz, the 16
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Micrel, Inc.
MIC2208
output impedance is dominated by the capacitance. A good approximation for transient response can be calculated from determining the frequency of the load step in amps per second;
f= A/sec 2
Then, determine the output impedance by looking at the output impedance vs frequency graph. Then calculating the voltage deviation times the load step; VOUT = IOUT x Z OUT The output impedance graph shows the relationship between supply voltage and output impedance. This is caused by the lower Rdson of the high side MOSFET and the increase in gain with increased supply voltages. This explains why higher supply voltages have better transient response.
a proper ring in tip measurement is required. Standard oscilloscope probes come with a grounding clip, or a long wire with an alligator clip. Unfortunately, for high frequency measurements, this ground clip can pick-up high frequency noise and erroneously inject it into the measured output ripple. The standard evaluation board accommodates a home made version by providing probe points for both the input and output supplies and their respective grounds. This requires the removing of the oscilloscope probe sheath and ground clip from a standard oscilloscope probe and wrapping a nonshielded bus wire around the oscilloscope probe. If there does not happen to be any non shielded bus wire immediately available, the leads from axial resistors will work. By maintaining the shortest possible ground lengths on the oscilloscope probe, true ripple measurements can be obtained.
Z TOTAL =
R DSON + DCR + X L GAIN
X COUT
Ripple measurements
To properly measure ripple on either input or output of a switching regulator,
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MIC2208
Package Information
12-Lead MLFTM (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http:/www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2005 Micrel, Incorporated.
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Revision History
Date 3-30-05 6-16-05 Edits by: Martin Galinski Martin Galinski Revision Number
September 2005
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